Saturday, 16 December 2017

Peerless Martin Taylor Maestro - a brief review




Peerless Martin Taylor Maestro

I got this from Lou at Guitars 'n Jazz in Summit, NJ, USA.

The guitar arrived on time from Fedex. After letting it acclimatize indoors for several hours, I opened the box. The string tension was lessened and paper plus packing lie between the strings and the fingerboard. Bits of packing and bubble wrap supported the guitar within its case. The case itself was sealed up and then wrapped in bubble wrap. Then, the cushioned guitar case was placed inside a stout cardboard box.  Got to give Lou and his crew 10/10 for packaging – I felt impressed by their packing job.

My first impression once holding this guitar was:  wow, is it ever small and light. At 15 inches wide and 2.75 inches deep, this guitar just cradles in your arms. It plays/feels very comfortably with a low fatigue factor. In fact, I can’t seem to put it down. No problem practicing >=3 hours per day on this axe.

The woodwork, binding and finish looks OK. This guitar plays like butter – unlike some of the vintage relic guitars I’ve tried over the years. It feels like playing a Les Paul with heavier gauge strings. It took me some time to mentally ingrain fret navigation as the fret markers are located on top and not in the fret space -- that coupled with the shorter than typical scale. The ebony fingerboard looks jet black. The frets seem well polished and I haven’t noticed any nut, fret or bridge buzz. The wide fret board feels very nice when going finger style.

With its solid top and small size, this guitar sounds bright acoustically. However, amplified, I’m able to get warm tones with no muddiness on the bass notes. You hear a scooped lower mid-range response compared to say an ES-175. The fundamental tones sound extremely clean and pure – pristine notes with a fast, woody attack – exactly why I wanted this guitar. 

Overall, I hear a brighter, resonant woody tone reminiscent of most solid top archies. I know some players prefer laminated top guitars for a subjectively more “wooden tone”. The solid versus plywood top debate proves endless. Get at least 1 of each might be the ultimate answer to that debate.

I played it with the stock round wound strings and then swapped in some new flat wounds strings since this is all I play on my arch tops. Fingerstyle provides a delicious tone with bass note clarity for days.  With a pick, this guitar also shines. I tried numerous picks and settled on the  D'Andrea Pro Plec 354 Shape (1.50mm). That pick with this guitar seem special. I played it through a 1963 black face Fender Deluxe reverb, my Polytone Megabrute and also a little Yamaha THH10 – it sounded good on all 3,

I  saw and heard few negatives. 1 concern: I had to tighten the nut holding 4 of the Grover tuners as they were just barely hand tight. I noticed this during my first string change. I also wish the pick guard was a little lower, however, its height gets constrained by the pickup and the volume control circuitry.

This guitar lacks a tone control circuit, but by lowering the volume pot, you roll off the highs enough that I don’t really miss a tone control. I also dislike a bridge pickup, switches, and 2 pairs of pots on my arch tops - so you now understand my bias.

The Kent Armstrong neck mounted pick up features Alnico 5 pole piece magnets and compliments the clear sonic tones emitted by the Maestro. . The ebony string trapeze contains  a wire to ground the strings. It's quiet plus free of hum and also surprisingly AF feedback resistant.

The Peerless Martin Taylor Maestro sounds lovely and plays beautifully --  if a 15 inch, shorter scale guitar is your cup of tea – it's worthy to consider.












Saturday, 9 December 2017

Op-amps Make Life Better

Op-amps thrill me.  I mostly employ them as instrumentation amps, plus AF small signal boosters or filters.  With simple math (often embedded in software or spreadsheets) , we can calculate design parameters such as gain, frequency response or better yet, a stage's transfer function.  Once built, you can measure and analyze your circuit's outcomes in your lab.

I get many ideas and information from peer-reviewed journals. If your affiliated with a university, then your librarian may have access to many free journal articles.  Occasionally, you must purchase a journal article if you deem it essential to your progress. This supports the author(s),  publisher and in turn, spawns more research and journal articles. Circle of life stuff in academia.

Article abstracts provide a good way to learn too.  I think of them as free little packets of information.  At the very least, they may inspire you down a new, exciting path. I read a lot of abstacts on various subjects.

Further, I try to devote some study time to instrumentation circuitry because you get exposed to some brilliant engineering techniques, historical perspectives and often enough, cool new parts.  Apart from the usual RF measurement gear, I hold a special interest in chemical and atmospheric measurement circuits.

OK, back to op-amps:

A case in point follows. Op-amps prove essential for modern instruments -- without them, our digital circuits often wouldn't have any usable voltage to work with.  Analog isn't dead Malcolm!

This abstract sums it up perfectly.

Abstract

The Analog Revolution and Its On-Going Role in Modern Analytical Measurements.

The electronic revolution in analytical instrumentation began when we first exceeded the two-digit resolution of panel meters and chart recorders and then took the first steps into automated control.

It started with the first uses of operational amplifiers (op amps) in the analog domain 20 years before the digital computer entered the analytical lab. Their application greatly increased both accuracy and precision in chemical measurement and they provided an elegant means for the electronic control of experimental quantities.

Later, laboratory and personal computers provided an unlimited readout resolution and enabled programmable control of instrument parameters as well as storage and computation of acquired data. However, digital computers did not replace the op amp's critical role of converting the analog sensor's output to a robust and accurate voltage.

Rather it added a new role: converting that voltage into a number. These analog operations are generally the limiting portions of our computerized instrumentation systems. Operational amplifier performance in gain, input current and resistance, offset voltage, and rise time have improved by a remarkable 3-4 orders of magnitude since their first implementations.

Each 10-fold improvement has opened the doors for the development of new techniques in all areas of chemical analysis. Along with some interesting history, the multiple roles op amps play in modern instrumentation are described along with a number of examples of new areas of analysis that have been enabled by their improvements.

Reference

Enke, C. G. (2015). The Analog Revolution and Its On-Going Role in Modern Analytical Measurements. Analytical Chemistry, 87(24), 11935-11947.


With new appreciation for op-amps from reading this article, I plan to get on the bench and have some fun applying them.  Who knows, I might blog some of these circuits 1 day.

Also, please continue to cherish and support science.  Evidence -- not hype nor hope should guide our daily decisions.

Saturday, 25 November 2017

Transistor Radio Series - The 7 MHz Scratch Synthesizer

Lab Notes from Transistor Radios


I like making simple component-level radios. This Fall, I rekindled my love for making transistor radios and hope to slowly blog some circuits and fun.  I'm warning you now — these circuits hearken the 1970s and 80s, SSD, EMRFD, old issues of Ham Radio and other stuff that require no coding skills.

For the Ham 40 meter band,  I thought about employing an Si5351 for the synthesizer -- naw. While this chip poses a great choice, I wanted something a little more organic. If you go the Si5351 route, I recommend you consider the offerings by EtherKit.  Jason toiled to improve the Arduino code library for this and perhaps you might support his future efforts?
 Above — Block diagram. This topology borrows from the concepts of Wes, W7ZOI.

[Section One — PLL Board]

Above —PLL Board Schematic. I built this whole project employing Ugly Construction.

Above — Reference oscillator output.  I built the 2 MHz crystal reference oscillator in a familiar Pierce circuit with an 74HC14. This hex inverter with Schmitt-trigger inputs makes a fabulous reference oscillator as seen above.


Above — DSO tracing of the 2 MHz reference divided by 13 in the 74HC193 4-bit synchronous counter. 
The internal 4-bit WORD gets programmed by pins 9, 10,1 and 15. By default, all 4 pins are set HIGH. Three N-channel MOSFETs locally switch the appropriate pins to ground according to a front panel 5-position switch.  This avoids the usual 4 toggle switches you might otherwise use.

For divide by 12, two steering diodes get employed without the 2K2 current-limiting resistors seen on the other switch positions. Each diodes' forward voltage drop limits the MOSFET gate drive current to ~ 1 mA which is about the current limit offered by the 2K2 resistor in the other switch positions.

The output of the 4-bit counter goes to Pin 3 of a phase-frequency detector built with a pair of 74HC74s.  The other 74HC74 clock input (Pin 11) is driven by a 74HC00 buffer which takes the low frequency, sine wave output of the offset mixer and squares it to proper CMOS voltage levels.

Above — The 74HC00 signal squarer in my DSO when tested with a 7.04 MHz signal generator.
With no AC input signal, a CMOS squarer may oscillate somewhere between between ~2 and 70 MHz.  My 5 volt 74HC00 DC supply is bypassed from AF to HF and the signal path input is also decoupled and bypassed with the 51 Ω / 470pF network.

For the PLL board, it's literally test as you go.  For example, build the  2 MHz Pierce oscillator and look at its output. Then use that output to test the 4-bit counter. Any old bench sine wave oscillator from AF to HF will test the 74HC00 squarer.  For the 2 remaining NAND gates: Ground the input pins of one gate while using the other gate to reset pins 1 and 13 in your phase/frequency detector.

I stuck with the classic, low-noise, OP-27 for the loop filter. You'll find the OP-27 in PLL circuits published decades ago. While we enjoy lower noise op-amps today, I got them for low cost long ago and they impart some nostalgia on my bench -- and the OP-27 is still a great part.

Since the VCO operates over a very narrow bandwidth [ hopefully 7.00 - 7.065 MHz ], you can filter the loop well. The 1K2 loop dampening resistor posed critical, so I employed a 1% part in that slot. The critical dampening resistance value in my loop was ~ 1170 to 1188 Ω. If I went below that resistance, the feedback loop goes into spasm and oscillates.  A 5% 1K2 resistor might not cut it!  Hence, a 1K5 Ω 5% part might be a better choice if you don't have a 1% 1K2 Ω resistor in stock.

For the op-amp filter, two, small, 63v, polyester 1 µF caps were placed in parallel since I lacked a 2.2 µF capacitor.  Above ~ 80 KHz, the op-amp loop filter = a third order filter.  At low frequencies it functions as a first order filter.


The filter allows the PLL to locks quickly and filters well. The output of the main VCO looks good for a home brew PLL system.
Above — The main VCO channel output (with some external attenuation) into my spectrum analyzer.  I could not see any reference oscillator spikes in the output.  Yay!  Basically, we're seeing the noise of my SA.

 [Section Two — VCO]

Above — The VCO schematic.  This VCO provides 3 output ports: main, offset mixer and a port to connect a frequency counter. A frequency counter proves essential, since this style of synthesizer uses a VXO with non-linear tuning;  plus frequency overlap or gaps may occur across different divisions of the reference oscillator.

The oscillator and 1 buffer run on a AF low-pass filtered 8 volt DC supply.  Any ripple on your oscillator's DC supply can pump the VCO and cause noise modulation.
Two outputs from the PNP Colpitts oscillator get lightly AC-coupled to an NPN buffer that I forgot to label.  Its 39 ohm collector resistor provides a low-amplitude signal which is further boosted by Q3 which drives a common base amp (Q4) to prevent any  of the frequency counter's digital noise from reaching the main oscillator.

The emitter of the unlabelled NPN buffer stage  functions as a normal emitter follower.  I ran 9 mA emitter current to preserve the main oscillator signal fidelity. This signal goes to an output port that connects to 1 of the offset mixer's input ports.
Above — The output of  Q4 ( frequency counter port) terminated in a 50 Ω  'scope input. When terminated with a high impedance it measured ~700mV Vpp. If you need higher peak to peak output voltage , run more current in the common base amp. Other ways to boost its output  include dropping the 470 Ω  DC voltage decoupling resistor to 150-220 Ω, and/or increasing the 470 pF coupling cap to 100 nF.

The main output circuitry takes signal tapped from the tank inductor L1. I tapped about 2/3 down and could have tapped it lower. In honesty, I was too lazy to rewind the L1 toroid, so I employed a 2.7p coupling cap to provide some low-level signal to Q5 without having to run high current to prevent further distortion of the VCO signal. A FET source current of 8.45 mA preserved the VCO signal fidelity and provides good drive for the final feedback amp.

My initial feedback amp transistor Q6 choice was a 2N2222A. I ran a target 23-24 mA to help achieve a stage output impedance of 50 Ω and raised the *51 Ω emitter resistor shown to 75 Ω to get this. Without a heat sink, the 2N2222a ran at 39-40 degrees C. Rather than make a heat sink, I stuck in a 2SC1971 and changed the emitter resistor to 51 Ω as shown in the schematic. I targeted ~23 mA emitter current.

This BJT runs at 28.7 degrees C and won't run away. The 2SC1971 I used was probably 1 of those cheap bootleg or counterfeit transistors as it cost < 40 cents with free shipping on eBay a few years ago.
I find these "probably bootleg" transistors work OK for HF projects. I've got some original,very old Mitsubishi 2SC1971s in my parts collection reserved for VHF amps + drivers and they cost a fortune now.


Above — A DSO capture of the unfiltered output from the 2SC1971 feedback amplifier (Q6) during some early experiments with attenuation on its output.


Above — An early DSO capture of the frequency counter output in yellow and the main VCO output in green with the low-pass filtered soldered in, but not the 6 dB pad.

Above — Final FBA version into my DSO.  I wanted an output close to 10 dBm so I can get ~7 dBm outputs from a quadrature hybrid coupler for use in single signal DC receivers if so desired.


Above — VCO board on my test bench. I soldered a 100K pot with 12.3 VDC supply, so that I could apply reverse DC voltage to the varactors and manually tune the VCO to test its function. Most importantly, this allows you to choose the correct tank inductor and fixed capacitor values.




Above — VCO board on my test bench.  As shown, I've got 4.86 volts manually applied to the varactors. I applied 3 back-to-back varactors, plus the small 47 p tank coupling capacitor to allow enough VCO tuning range with the lowest pk-pk AC voltage possible on the varactors to potentially reduce VCO phase noise.

When you connect the varactors to the VCO, tweak the small 5-30p trimmer cap to get phase lock. Switch off the power. Then after 4-5 seconds, switch back on the12 VDC. If your VCO does not phase lock, tweak the trimmer cap some more and repeat. You can connect your voltmeter right to the PLL output to monitor the DC voltage.

My circuit captures and locks the VCO after switching it off and then on with 3.9 volts DC measured on the PLL output with the VXO set to its lowest frequency + the 4-bit counter set to divide by 11. It's then set and forget.  Once set, the VCO locks perfectly every time and it stays on the set frequency for days.

 [Section Three — Offset Mixer]

Above — Offset mixer schematic. 

In old CB radio synthesizers, engineers often used a single BJT as an offset mixer (usually they ran 1 for TX and another 1 for RX). To prevent mixer output from leaking out its AC and DC ports, we may do 2 things: well filter the DC going to the BJT -- plus run common-base amps on the 2 BJT mixer inputs to provide reverse isolation for the mixer board input ports.

The VCO and VXO ports get the same common-base amp. An active low-pass (ripple filter), plus some serious decoupling + HF to VHF bypass filter the DC power line. The mixer output gets a simple, but stout, pi low-pass filter. The shunt 560 Ω resistor loads the output to stabilize the output signal + boost filtration.



Above — DSO capture with 10X probe at the output of the 1 µF capacitor. At this point the only filtration is provided by the 1n capacitor shunt to ground, so you still see some mixer products.


Above —  DSO capture with 10X probe at the output of the pi low-pass filter. You've filtered it enough when it looks like a sine wave at the correct offset output frequency. I tested it with the project VXO and a bench signal generator.

 [Section Four — VXO]

We require a VXO that tunes the VCO from 7.0 to 7.065 MHz. I had some HC49-S style 28.4 MHz crystals and hoped to divided them by 4 -- and get enough frequency swing to tune my desired VCO range. I failed here!  The crystals on hand proved terrible and would not pull as far as I'd hoped without compromising my desired less than 1 Hertz VXO temperature drift fluctuation. 

I tried them in a super VXO fashion + varied the inductance and such but could not get a fully meshed to fully open tuning capacitor swing over 10 KHz with a divide by 4.  In my collection, I found an air variable capacitor that offered more range, but alas it suffered backlash and when I mounted it in my enclosure the Bakelite support material cracked and crumbled rendering it garbage.

I hope to get some different 28.8 MHz crystals and derive the tuning range I initially calculated and hoped to achieve. I will also try to find an air variable capacitor with a lower minimum capacitance like the 1 that broke. Perhaps it was foolhardy to chose a divide by 4 scheme? We'll see.

Above —  VXO schematic. The single hex inverter buffer might not be needed, however, I've got lots of CMOS logic to use up and more years behind me than ahead.

Final synthesizer tuning range  = 7.007 to 7.065 MHz with some gaps as follows:


The VXO tunes from 7.189152 to 7.19702 MHz. As mentioned, Ill work on the VXO to remove the gaps in ranges 1, 2 and 3 -- and hopefully get down close to 7.00 MHz.
Limiting the VXO range to only that what's needed improves the synthesizer tuning resolution.

 [Section Five — Photos]


Above —  All 4 boards were built in re-purposed Hammond boxes. A PIC-based counter sits on top of the offset mixer. I build modular gear and this allows modification and fosters experimentation. When I build a final transistor radio receiver, I plan to place the offset mixer, PLL circuitry and VXO on the same board inside the radio with some shielding. My VCOs always go in a RF tight container. A 0.0033 µF feed through capacitor connects the VCO varactors to the outside world.

Making this project I learned much and like my VCO + mixer, the sublime loop filter and also avoiding toggle switches by running local MOSFET switches. I've already extended the MOSFET switching technique in a more complex PLL for a scratch UHF synthesizer.

 Above —  All 4  boards on my bench

Above —  Main VCO output in a DSO.

[Section Six — Miscellany]
Above —  Basic configuration of the HC193 4-bit synchronous counter. By setting the 4 switches HIGH or LOW you can divide by 1 to 15.



Above — Cascading two 4-bit counters yields division by 1 to 255. If you were to place all 8 switches on a front panel, we would place the left counter's switches to the right of the 240 weight switches so that we could program it in a more "humanly logical" fashion.

It's great fun to play with old school digital circuitry such as these CMOS chips. I've got the whole series of HC74 synchronous and asynchronous counters in my parts bins.

Above — 3 edge-triggered phase and frequency detectors using D flip flop(s). I normally employ Figure A with active filtering. The op-amp filter boosts the DC output signal up in voltage which may help improve VCO phase noise and stability.

Figure B employs a charge pump. You'll see diodes or transistors used in charge pumps -- often in PLL circuits within ICOM, Yaesu and other brand-name radios.

Compared to the Exclusive OR phase detector ( in the 4046 etc.), edge-triggered PD's exhibit a greater linear tuning range, plus better capture, lock and tracking characteristics. All of them may be effected by input signal duty cycle. 50% proves best.